-
2022-09-23 11:14:56
AD694 is a monolithic current transmitter
feature
4–20 mA, 0–20 mA output ranges Pre-calibrated input ranges: 0V to 2V, 0V to 10V; precision voltage reference; programmable to 2.000 V or 10.000 V; single-supply or dual-supply operation; Wide supply range: 4.5 V to 36 V; wide output compliance; input buffer amplifier; open loop alarm; optional external pass transistor; self-heating error; 0.002% typical nonlinearity.
Product Description
The AD694 is a monolithic current transmitter that accepts a high-level signal input and drives a standard 4-20 mA current loop used to control valves, actuators, and other devices commonly used in process control. The input signal is buffered by an input amplifier that can be used to scale the input signal or buffer the output of a current-mode DAC. Pre-calibrated input ranges of 0 V to 2 V and 0 V to 10 V are selected with simple pin strapping; other ranges are programmable using external resistors.
Output stage compliance extends to within the 2V range of VS, and its special design allows the output voltage to extend below the common voltage in dual supply operation. Alarm warns of 4–20 mA open circuit or inconsistent output stage.
Active laser trimming of AD694 thin film resistors achieves high accuracy without the need for additional tuning and calibration. External pass transistors can be used with the AD694 to reduce load power dissipation and expand the operating temperature range.
The AD694 is an ideal building block for systems requiring noise-resistant 4-20 mA signal transmission to operate valves, actuators, and other controls, and for transmitting process parameters such as pressure, temperature, or flow. It is recommended as an alternative to discrete designs in various applications in industrial process control, factory automation and system monitoring.
The AD694 is available in hermetic, 16-pin CERDIP and plastic SOICs specified over the industrial temperature range of -40°C to +85°C and in 16-pin plastic DIPs specified at 0°C to +70°C used within the temperature range.
Product Highlights
1. AD694 is a complete voltage input to 4–20 mA output current transmitter.
2. Pin programmable input ranges are pre-calibrated to 0 V to 2 V and 0 V to 10 V.
3. The input amplifier can be configured to buffer and scale the input voltage, or be used as an output amplifier for the current output dac.
4. The output voltage compliance extends to within 2V of the positive power supply, which is lower than the public power supply. When working with 5V power supply, the consistency of output voltage is 30V lower than that of ordinary power supply.
5. The AD694 interfaces directly with 8-, 10-, and 12-bit single-supply CMOS and bipolar DACs.
6. 4 mA zero current can be turned on and off via aTTL control pins, allowing 0-20 mA operation.
7. Open collector alarm warns of loop failure due to open circuit or inconsistent output stage.
8. The provided monitor output drives an external transistor. It features no-load power consumption, expanded operating temperature range, and reduced self-heating errors.
Function description
The operation of the AD694 can be understood by dividing the circuit into three functional parts (see Figure 1). First, a single-supply input amplifier buffers a high-level single-ended input signal. The buffer amplifier drives the second part, the voltage-to-current (V/I) converter, which produces a signal-dependent current of 0 to 16 mA.
The third part is the voltage reference and offset generator, which is responsible for providing the 4ma offset current signal.
buffer amplifier
The buffer amplifier is a single-supply amplifier that can be used as a unity-gain buffer, a current output amplifier with maximum RL and a supply voltage output DAC, or as a gain block to amplify low-level signals. The PNP input stage of this amplifier has a common-mode range that extends from a few hundred millivolts below ground to 2.5 volts, compared to the amplifier's Class A output which appears at pin 1 (FB). When the amplifier operates as a follower, the output range extends from about 1 millivolt above common to within 2.5 volts of VS. The amplifier can supply a maximum load of 5 K, but can only sink as much as its internal 10 KΩ pull-down resistor allows.
V/I Converter The ground-referenced input signal from the buffer amplifier is converted through A2 to a current of 0 to 0.8 mA, level shifted to the positive supply. The current mirror then multiplies this signal by a factor of 20, making the signal current from 0 to 16 mA. This technique allows the output stage to drive the load to within 2V of the positive supply voltage referenced power supply rejection (VS) amplifier A2 passes the voltage at pin 1 across resistors R1 and R2 by driving Darlington transistor Q2. The high gain Darlington delivers resistive current to its collector and R3 ( 900 Ω). A3 forces a current gain of 20 by level-shifting the signal through a 45Ω resistor.
Therefore, the transfer function of the V/I stage is:
resulting in a 0-16 mA output swing from a 0-10 V input. Connecting pin 4 (2 V FS) to ground shorts R2, resulting in a 2 V full-scale input with a 16 mA output range.
The output stage of the V/I converter is uniquely designed to allow the IOUT pin to drive loads below the device common (substrate) potential. The output transistor can always drive the load to a point 36 V below the positive supply (VS). One IOUT: Voltage vs. Temperature Compliance An optional NPN pass transistor can move most of the power dissipation off-chip, extending the operating temperature range.
The output stage is current limited to approximately 38mA to protect the output from overdriving at its input. V/I will allow linear operation to about 24mA. The V/I converter also has an open collector alarm (pin 10) which warns of an open circuit condition on the input pins or an attempt to drive the output above VS – 2 V.
4ma offset generator
This circuit converts a constant voltage from a voltage reference to a constant current of approximately 200 μA. This current is summed with the signal current at pin 14 (BW regulation), resulting in a constant 4mA offset current in IOUT. The 4mA trim (pin 6) allows the offset current to be trimmed to any current in the range of 2mA to 4.8mA. Pin 9 (4mA on/off) can completely turn off the bias current, allowing AD694 operation from 0 to 20mA if it is boosted to 3.0V or more. In normal 4–20mA operation, pin 9 is grounded.
voltage reference
The user can use a 2V or 10V voltage reference, selectable via pin strapping. The 10V option is available for supply voltages greater than 12.5V, and the 2V output is available for the entire 4.5V to 36V supply range. This reference can supply up to 5 mA for user applications. A boost transistor can be added to increase the current drive capability in 2V mode.
Apply AD694
The AD694 can be easily connected to dual-supply or single-supply operation to operate from supplies as low as 4.5 V and as high as 36 V. The following sections describe different connection configurations and how to adjust them. Table 1 shows possible connection options.
Basic connections: 12.5 V single supply operation, 10 V FS
Figure 2 shows the minimum connections required for basic operation with a 12.5 V supply, 10 V input range, 4–20 mA output range, and 10 V voltage reference. The buffer amplifier is connected as a voltage follower to drive the V/I converter by connecting FB (pin 1) to –Sig (pin 2). The 4mA on/off (pin 9) is tied to ground (pin 5) to enable the 4mA offset current. The AD692 can drive a maximum load of RL = [VS - 2 V] / 20 mA, so the maximum load is 525 Ω for a 12.5 V supply.
Select 2 V full-scale input
The 2V full-scale option is selected by shorting pin 4 (2V FS) to pin 5 (common). The connections should be as short as possible; any parasitic resistance will affect the pre-calibrated range accuracy.
Set the voltage reference to 2 V output by shorting pin 7 to pin 8 (10 V force to 2 V sensing). If desired, a 2V reference can be set for the remote force and sensing connections. Keep in mind that the 2V sense line has a constant current of 100 microamps, which can cause offset errors on long lines. A 2V reference voltage option is available for all supply voltages greater than 4.5V.
An NPN boost transistor can be added in 2v mode to increase the current drive capability of the 2v reference. The 10V force pin is connected to the base of the NPN and the NPN transmitter is connected to the 2V sense pin. The minimum Vs of the part increases by about 0.7 V.
4.5V Single Supply Operation
For operation with a 4.5 V supply, the input span and voltage reference output must be reduced to provide the amplifier with the required 2.5 V operating headroom. This is done by adjusting the AD694's 2 volt full-scale input and voltage reference output to 2 volts as described above.
General Design Guidelines
A 0.1µF decoupling capacitor is recommended in all applications from VS (Pin 13) to Com (Pin 5). If the output load is non-resistive, additional parts may be required, see section Driving non-resistive loads. The negative voltages at the PNP inputs of the buffer amplifiers should not exceed -0.3v, otherwise they will start to draw a lot of current. If there is a danger of this happening, an input protection resistor must be added to the input. The output pin 1 (FB) of the buffer amplifier is not short-circuit protected. If you short this pin to ground or if it corresponds to a signal present on the amplifier, you can damage this pin. The input signal should not drive pin 1 (FB) directly; always use a buffer amplifier to buffer the input signal.
Driving non-resisting loads
The AD694 is stable when driving resistive loads. As shown in Figure 3, adding a 0.01-µF capacitor between IOUT (Pin 11) and Com (Pin 5) ensures the stability of the AD694 when driving inductive or poorly defined loads. This capacitor is recommended when there is any uncertainty in the load characteristics.
Additional protection is recommended when driving inductive loads. Figure 3 shows two protection diodes, D1 and D2, used to prevent voltage spikes above VS or below common voltage spikes that could damage the AD694. These diodes should be used in addition to the 0.01µF capacitor. When using the optional NPN transistor, the capacitor and diode should be connected to the NPN emitter, not to Pin 11.
0-20mA operation
The AD694 has an output range of 0-20mA and eliminates the 4mA bias current by using the 4mA on/off pin. In normal 4–20mA operation, the 4mA on/off (pin 9) is tied to ground, enabling the 4mA bias current. Connecting pin 9 to a potential of 3V or greater turns off the 4MA bias current; connecting pin 9 to a 10V reference, positive supply, or a TTL control pin is a convenient method. In 0–20 mA mode, the input span is increased by 20%, so the pre-calibrated input spans of 2 V and 10 V become 2.5 V and 12.5 V. The minimum supply voltage for both spans is increased to 5 V and 15 V.
The 4 mA on/off pin can also be used as a "rocker pin" to loosen a valve or actuator, or as a way to completely close a 4-20 mA loop. Note that the pin only removes the 4mA offset, not the signal current.
Dual Power Operation
Figure 4 shows the AD694 operating in dual-supply mode. (Note that a pass transistor is shown, and dual-supply operation is not required.) The device is powered entirely from a positive supply as low as 4.5 V. The unique design of the output stage allows the output pins to extend below the common line of the negative supply. The output stage delivers current to the 36 V point below the positive supply. For example, when operating from a 12.5 V supply, the AD694 can source current to points 23.5 V below the common supply. This feature simplifies interfacing with dual-supply DACs, eliminating grounding and level-shift issues, while increasing the load the transmitter can drive. Note that the IOUT pin is the only pin that is allowed to extend below -0.3 V common.
Operates with pass transistors
The AD694 can operate as a standalone 4–20 mA converter without additional active components. However, provision has been made to connect IOUT to the base of an external NPN pass transistor as shown in Figure 4. This allows most of the power dissipation to be moved out of the chip to improve performance and extend the operating temperature range. Note that the positive output voltage compliance is reduced by approximately 0.7 V, the VBE through the device. When the AD694 operates with dual power supplies, as shown in Figure 4. This does not degrade the voltage compliance of the output stage.
The selected external pass transistor should have a bveceo larger than the expected supply voltage and sufficient power rating to operate continuously at 25mA at the supply voltage. FT should be in the range of 10 MHz to 100 MHz, and β should be greater than 10 at 20 mA emitter current. An external pass transistor heat sink method is recommended.
Power Considerations
The AD694 is rated to operate over its specified temperature without the use of an external pass transistor. However, it is possible to exceed the absolute maximum power dissipation with some combination of supply voltage and voltage reference load. The internal dissipation of the section can be calculated to determine if it is possible to exceed the absolute maximum dissipation. The mold temperature must not exceed 150°C.
The total power dissipation (PTOT) is the sum of the power dissipated by the internal amplifier P (static), the voltage reference P (VREF) and the current output stage P (IOUT) as follows:
PTOT = P(Standing) + P(VREF) + P(IOUT)
in:
P (Standing) = 2 mA (max) × VS;
P(VREF) = (VS – VREF) × IVREF;
P(IOUT) (VS – VOUT) × IOUT (max):
IOUT (max) can be the maximum expected operating current or overdrive current of the device.
P(IOUT) falls to (2V×IOUT) if a pass transistor is used.
definition:
VREF=reference output voltage
IVREF=reference output current
VS = supply voltage
VOUT = voltage at the output pin.
An appropriate safety factor should be incorporated into the PTOT.
The junction temperature can be calculated using the following formula:
TJ = PTOT (θJC + θCA) + TAMBIENT
T-type J=P total number (θθθ is the thermal resistance between the chip and the package (case), θ is the thermal resistance between the case and its surrounding environment, which is determined by the thermal connection characteristics of the case and the surrounding environment.
For example, assume the part is operating at 50°C with a VS of 24 V in a CERDIP package and a load of 1 mA on a 10 V reference. Assuming IOUT is grounded, the maximum IOUT is 20 mA. Internal dissipation is:
P(TOT) = 2 mA × 24 V + (24 V – 10 V) × 1 mA + (24 V – 0 V) × 20 mA
= 48 mW + 14 mW + 480 mW = 542 mW
Using a 30°C/W theta and a 70°C/W theta (from the spec page), the junction temperature is: JC Corporation California
TJ = 542 mW (30°C/W + 70°C/W) + 50°C = 104.2°C
Internal power dissipation can be reduced by reducing the value of θ by using airflow or a heat sink, or by reducing the PTOT of the AD694 by using an external pass transistor. Figure 5 shows the maximum case and still air temperature for a given power level.
Adjustment Program
The following sections describe methods for adjusting the output current offset, span, and voltage reference.
Adjust 4 mA zero point
The 4ma zero current can be adjusted from 2ma to 4.8ma to accommodate large input signal excursions, or to allow small adjustments in zero current. The zero point can be adjusted by pulling pin 6 up or down (4 mA adjustment) to increase or decrease the nominal offset current. 4 mA adjustment. (Pin 6) should not be driven to voltages greater than 1 V. The arrangement of Figure 6 will give an approximately linear adjustment of the 4 mA offset within fixed limits. To find the proper resistor value, first choose X, the desired adjustment range is a fraction of 4 mA. Substitute this value with the selected reference output voltage (usually VREF = 2V or 10V) with the appropriate formula below to determine the required resistor value.
RP=180 8486 ; (1/X–4.5)
RF=500Ω[(VREF/1.22V)–0.18–0.82X][1/X–4.5]
These equations take into account the ±10% internal resistance tolerance and ensure a minimum adjustment range of 4 mA offset. For example, assume the 2 V reference option has been selected. Choosing X=0.05 gives an adjustment range of ±5% of the 4 mA offset.
RP=180ΩΩ(1/0.05–4.5)=2.79k
RF=500Ω×[(2V/1.22)–0.18–0.820.05][1/0.05–4.5]=10.99kΩ
These values can be rounded to more convenient values of 2.5 kΩ and 9.76 kΩ. In general, if the value of RP is rounded down slightly, the value of RF should be rounded down proportionally, and vice versa. This helps keep the adjustment range symmetrical.
Adjustment range of 10 V FS
When the AD694 is configured with a 10 V input full scale, the network shown in Figure 7 can be used to adjust the range. This scheme allows for approximately linear adjustment of the span above or below the nominal value. Span adjustment does not interact with the 4 mA offset.
To select RS and RT, select X, the desired adjustment range, as part of the range. Replace this value in the appropriate formula below.
RT = 1.8 kΩ((1 – X)/X)
RS = 9 kΩ[1 – 0.2 (1 + X)( 1 – X )] / 2X
These formulas take into account the ±10% absolute resistance tolerance of the internal span resistors and ensure a minimum span adjustment range. For example, choosing an adjustment range of ±2% or 0.02 gives:
RT = 1.8 kΩΩ((1 – 0.02) / 0.02) = 88.2 k.
RS = 9 kΩ×[1 – 0.2 (1 + 0.02)( 1 – 0.02 )] / (2 0.02) = 175.5 kΩ
These values can be rounded to the more convenient values of 100 kΩ and 198 kΩ. In general, if RT is rounded, then the value of RS should be rounded proportionally, and vice versa.
Adjustment range of 2 V FS
Due to the single-supply nature of the AD694, the pre-calibrated 2V full-scale range requires a different adjustment scheme. Figure 8 shows an adjustment scheme that allows for an approximately linear adjustment of the 2 V span roughly plus or minus the nominal value. Span adjustment does not affect the value of the 4 mA bias current.
To find the proper resistor value, first select X, the desired adjustment range is part of the output range. Substitute this value into the following formula:
RA= 2 × X × RB where RB is greater than 5 K
RC = (2.75 kΩ×1 – 0.275X) X)/(These formulas take into account the ±10% absolute tolerance of the interspan resistors and ensure a minimum adjustment range.)
For example, choose the adjustment range as ±320μA of FS or, ±2%, set X=0.02. therefore:
Set RB=10K, then RA=2(.02)×10KΩ=400ΩRC=(2.75KΩ×0.02)/(1–0.275×(0.02))=55.3Ω
The value of RC can be rounded to a more convenient value of 49.9Ω. In general, if RA is rounded, then RC should be rounded proportionally, and vice versa; rounding increases the adjustment range.
Plan other spans
There are two ways to program input spans less than 10V. The first method reduces the input span by programming an irreversible gain in the buffer amplifier. For example, to achieve an input span of 0-5v, the AD694 is set to its 10v full-scale mode, and by adding 2 resistors, the buffer amplifier is configured for an irreversible gain of 2. Now, a 5v signal at +Sig produces a 10v full-scale signal (Pin 1) at FB, which is the input to V/I. This method requires V/I to be programmed to 10v full scale for input ranges between 2v and 10v. If an input range less than 2v is required, it should be programmed to 2v full scale. This adjustment scheme makes the accuracy of the span adjustment dependent on the ratio accuracy of the desired gain resistors. Therefore, ranges other than 2V or 10V can be precisely configured without the use of trimmer potentiometers, provided that the resistance ratios are accurate enough. A span between 2 V and 10 V requires a supply voltage of 12.5 V. Spans below 2 V require a voltage ratio of 4.5 V or higher.
The second is to allow programming of other spans less than 10 V when the supply voltage is less than 12.5 V. Since the AD694 amplifier requires 2.5 V of headroom to operate, a 7.5 V supply can be used for a 5 V full-scale input. This is accomplished by placing a resistor in parallel with R2 (2v FS[Pin 4] to Com[Pin 5]) to adjust the transconductance of the V/I converter without incurring a headroom penalty. The downside of this approach is that the external resistors must match the internal resistors exactly, so scaling is required. When choosing this value, the ±10% uncertainty of the absolute value of internal resistor R2 should be considered.
Adjust the reference output
Figure 9 shows a way to make a small adjustment to the 10V reference output. The circuit allows a linear adjustment range of ±200 mV. The 2V reference can also be adjusted, but only in the positive direction.
Other reference voltages can be programmed by adding external resistors. For example, a resistor in parallel with R5 can be added to boost the reference output to 20 V. Instead, the reference voltage can be set to a value between 2 V and 10 V using a resistor in parallel with R6. The output voltage VREF=2 V(R6+R5)/R5. When choosing an external trim resistor, keep in mind that the absolute resistance tolerance of the internal resistor is only ±10%, while the ratio is matched to high precision. Be prepared to compensate for this if you need an exact voltage other than the pre-calibrated value of 2 V or 10 V.
Bandwidth Control
The bandwidth of the AD694 can be limited to provide noise filtering. This is achieved by connecting an external capacitor from BW ADJ (pin 14) to VS (pin 13) as shown in Figure 10. To program the bandwidth, substitute the desired bandwidth (in Hz) into the following formula to determine the required capacitor.
C=1/(2π××900Ω) BW
Due to internal resistor tolerances, the selected bandwidth will vary by ±10%, plus an additional amount due to capacitor tolerances.
This method of bandwidth control is not recommended for filtering large high frequency transients in the input signal. It is recommended to use an input filter to remove frequencies greater than the BW of the buffer amplifier to avoid input amplifier correction noise.
Buffer Amplifier Offset Adjustment
The buffer amplifier input voltage offset has been laser trimmed to high precision; however, in some cases, offset trimming may be required. Figure 11 shows the trimming method; the trimming range for this scheme is greater than ±2.5 mV. This trimming method is not recommended to affect the 4mA bias current, as trimming will cause bias drift into the buffer amplifier. The buffer amplifier will drift about 1 volt/°C for every 300 volts of inductive bias. To adjust the 4mA offset current, see the Adjusting the 4mA Zero Point section.
Alarm circuit
The AD694 has an alarm circuit that warns of an open circuit condition at IOUT (pin 11), or an attempt to drive IOUT above VS–2 V. If a runaway condition is detected, the alarm transistor will pull down. The alarm current is limited to around 20mA.
Figure 12 shows a typical application. In digital/analog systems, the alarm can provide a TTL signal to the controller. The collector of the alarm transistor is connected to the system logic power supply through a 20 kΩ pull-up resistor. During normal operation the alarm is turned off and the alarm pin voltage is high. The ALERT pin is driven low if the wire to IOUT (pin 11) is broken, or if a large input overdrive forces IOUT too close to VS. This configuration is compatible with CMOS or TTL logic levels. Alarm transistors can also be used to directly drive LEDs or other indicators.
application
Current output DAC interface
The AD694 can easily be connected to a current output DAC, such as the AD566A, to build a digital to 4–20 mA interface, as shown in Figure 13. The AD694 provides the voltage reference and buffer amplifiers required to operate the DAC. Constructing a circuit requires only simple connections.
The AD694's 10V reference provides AD 566. The buffer amplifier converts the full-scale current to +10 V using the internal resistors in the DAC; therefore the AD694 is configured as a 10 V full-scale input. The 10pf capacitor compensates for the DAC's 25pf output capacitance. An optional 100Ω trimmer resistor (RT) allows full-scale trimming, the 50Ω resistor can be replaced if trimming is not required; accuracy is typically ±1 LSB, trimming does not affect the 4mA offset. Care should be taken when managing circuit grounding. The AD694 pins 9, 3 and AD566 pins 3 and 7 should be connected as short as possible and connected to a point near pin 5 of the AD694. Best practice is to connect from each pin individually to star ground; this is required for the AD566 power ground, from pins 12. The 4–20 mA output (pin 11) must have a return to power ground. Depending on the size of the load to be driven and power dissipation considerations, the load's return line can be connected to either power ground or a -15 V supply.
A 12-bit input of a 4-20 mA output interface can operate on a 15-volt supply. The DAC operates in its voltage switching mode; this allows the DAC to provide an output voltage proportional to the digital input code, ranging from 0v to VREF, while providing a voltage reference less than 2.5v. The AD694 voltage reference is connected to a supply of 2V, and the input stage is set to 2V full-scale; the input buffer amplifier is used to buffer the voltage at the DAC output. Connected this way, a full-scale DAC input code will result in a 20mA output, while an all-zeros code will result in a 4mA output. The load of the AD694 voltage reference is code dependent, and the response time of the circuit will be determined by the response of the voltage reference. The supply voltage to the AD7541A should be kept around 15V. If VS is significantly reduced from 15V, the differential nonlinearity of the DAC will increase and the linearity will decrease.
In some applications, at 4-20 mA output, some underrange and overrange are required. For example, assuming an overrange and underrange capability of ±5% of range is desired, the output current range corresponding to the full scale of the DAC is 3.2ma to 20.8ma. To do this, the span of the AD694 will be increased by 10% to 17.6 mA by adding a non-inverting gain of 1.1 in the buffer amplifier. Then, by using the adjustment scheme explained in the Adjusting the 4ma Zero section, the 4ma offset will be reduced by 0.8ma. Then a digital input from all zeros to full scale will result in an output current of 3.2mA to 20.8mA.
low cost sensor transmitter
Sensor bridges typically output differential signals in the full-scale range of 10 mV to 100 mV. Using the AD694, dual op amps, and some resistors, an instrumentation amplifier front end can be added to easily handle these types of low-level signals.
The traditional 3-op amp instrumentation amplifier uses the AD708 dual-op amp front end, and the AD694 buffer amplifier is used for the subtractor circuit, as shown in Figure 15. The AD694's 2 volt reference is used to provide 2 volts of "ground" to ensure proper operation of the in-amp over a wide common mode range. The reference pin of the subtractor circuit is connected to the 2V reference (point C). A 2 kΩ pull-down resistor ensures that the voltage reference can sink any subtractor current. 2V FS (pin 4) is connected to the 2V reference; this will offset the V/I converter's input range by 2V positive to match the "ground" of the input amplifier.
The AD694 will now output an output current of 4–20 mA for a 0 V to 2 V trans-VA differential swing. Adjust the gain of the front end of the input amplifier to obtain the desired full-scale input signal at the VIN, resulting in a VA of 2 V. For example, a sensor with a full scale of 100 millivolts would require a gain of 20 on the front end. The gain is determined according to the following equation:
G = [2RS/Rg] + 1
The circuit shown will draw VINs at 4–20 mA current. The common mode range of this circuit is 3v to 8v. The low end of the common-mode range is limited by the AD708's ability to pull down on RS. The common mode range can be extended to around 1.5v using a single supply amplifier.
As shown, the circuit handles positive differential signals (VIN positive). To handle bipolar differential signals (VIN positive or negative), the reference pin of the input amplifier (point C) must be positively offset from the 2V reference. For example, disconnecting point C from the 2V reference and connecting it to a 3V supply will result in a VA of 1V (or half-scale) for the sensor's zero-volt differential input.