feature

4.5V to 16V operating input voltage range; high 32 mΩ internal PFET switch; efficiency: up to 95%; internal soft-start; output voltage adjustable to 0.8V; 5A continuous output current; fixed 500kHz PWM operation; cycle-by-cycle current limit ; short circuit protection; thermal shutdown; small SO-8 and DFN-8 packages.

application

Point-of-load dc/dc conversion; PCIe graphics cards; set-top boxes; DVD drives and HDDs; LCD panels; cable modems; telecom/networking/datacom equipment.

General Instructions

The AOZ1014 is an efficient, easy-to-use 5A buck regulator. The AOZ1014 operates from a 4.5V to 16V input voltage range and provides up to 5A continuous output

The output voltage is adjustable down to 0.8V of current. The AOZ1014 is available in SO-8 and DFN-8 packages and is rated for an ambient temperature range between -40°C to +85°C.

typical application

Detailed description

The AOZ1014 is a current mode buck regulator that integrates a high-side PMOS switch and a low-side free-wheeling Schottky diode. It operates over an input voltage range of 4.5V to 16V and provides up to 5A of load current. The duty cycle can be adjusted from 6% to 100%, allowing a wide range of output voltages. Features include enable control, power-on reset, input undervoltage lockout, fixed internal soft-start, and thermal shutdown.

The AOZ1014 is available in SO-8 and thermally enhanced DFN-8 packages.

Enable and Soft Start

The AOZ1014 has an internal soft-start function to limit inrush current and ensure a smooth rise of the output voltage to the regulated voltage. The soft-start process begins when the input voltage rises to 4.0V and the voltage on the EN pin is high. During the soft-start process, the output voltage gradually rises to the regulated voltage within typically 4ms.

4ms soft-start time is set internally.

The EN pin of the AOZ1014 is active high. If the enable function is not used, connect the EN pin to the VIN. Pulling it to the ground will disable the AOZ1014. Don't leave it on. The voltage on the EN pin must be higher than 2.0 V to enable the AOZ1014. When the voltage on the EN pin falls below 0.6 V, the AOZ1014 is disabled. If the application circuit requires that the AOZ1014 be disabled, an open drain or open collector circuit should be used to connect to the EN pin.

steady state operation

Under steady-state conditions, the converter operates in fixed frequency and continuous conduction mode (CCM).

The AOZ1014 integrates an internal P-MOSFET as a high-side switch. The inductor current is sensed by amplifying the voltage drop from the drain to the source of the high-side power MOSFET. The output voltage is reduced by an external voltage divider at the FB pin. The difference between the FB pin voltage and the reference voltage is amplified by an internal transconductance error amplifier. At the PWM comparator input, the error voltage displayed on the COMP pin is compared to the current signal that is the sum of the inductor current signal and the slope compensation signal. If the current signal is less than the error voltage, the internal high side switch is turned on. Inductor current flows from the input through the inductor to the output. When the current signal exceeds the error voltage, the high side switch is turned off. The inductor current is free-spinning output through an external Schottky diode.

The AOZ1014 uses a P-channel MOSFET as the high-side switch. It saves the bootstrap capacitance typically seen in circuits using NMOS switches. The upper switch is allowed to be turned on 100%, and the linear adjustment operation mode is realized. The minimum voltage drop from V to V is the load current times the DC resistance of the MOSFET plus the DC resistance of the buck inductor. Its calculation formula is as follows:

where VoxMAX is the maximum output voltage; VIN is the input voltage between 4.5V and 16V; IO is the output current from 0A to 5A; RDS(ON) is the internal on-resistance MOSFET with a value between 25mΩ and 55mΩ, depending on on input voltage and junction temperature; inductor DC resistance.

On-off level

The AOZ1014 switching frequency is fixed and set by the internal oscillator. Due to device variations, the actual switching frequency can range from 350kHz to 600kHz.

Output voltage programming

The output voltage can be set by feeding back the output to the FB pin and a resistor divider network. in the application circuit shown in Figure 1. The resistor divider network consists of R and R. Typically, a design is started by picking a fixed value of R and calculating the required R1 using the formula below.

Table 1 lists some standard values R, R for the most commonly used output voltage values.

The combination of R1 and R2 should be large enough to avoid drawing too much current from the output, which will cause power loss.

Since the switch duty cycle can be as high as 100%, the maximum output voltage can be set to the high input voltage minus the voltage drop across the upper PMOS and inductor.

Protection features

AOZ1014 has multiple protection functions to prevent damage to the system circuit under abnormal conditions.

Over Current Protection (OCP)

The sensed inductor current signal is also used for overcurrent protection. Since the AOZ1014 uses peak current mode control, the COMP pin voltage is proportional to the peak inductor current. The COMP pin voltage is internally limited between 0.4V and 2.5V. The peak current of the inductor is the automatic limit cycle.

The cycle-by-cycle current limit threshold is set between 6A and 8A. When the load current reaches the current limit threshold, the cycle-by-cycle current limit circuit immediately turns off the high-side switch to terminate the current duty cycle. The inductor current stops rising. Cycle-by-cycle current limit protection directly limits the inductor peak current. Due to the limitation of peak inductor current, the average inductor current is also limited. When the cycle-by-cycle current limit circuit is triggered, the output voltage drops as the duty cycle decreases.

The AOZ1014 has internal short-circuit protection to prevent catastrophic failure under output short-circuit conditions. The FB pin voltage is proportional to the output voltage. When the FB pin voltage is lower than 0.2V, the short circuit protection circuit is triggered. As a result, the converter is turned off and hiccups at a frequency equal to 1/8 of the normal switching frequency. Once the short-circuit condition disappears, the drive will start with a soft start. In short-circuit protection mode, the average current of the inductor is greatly reduced due to its low disturbance frequency.

Power-On Reset (POR)

A power-on reset circuit monitors the input voltage. When the input voltage exceeds 4V, the inverter starts to work. When the input voltage drops below 3.7V, the inverter will shut down.

Thermal Protection

An internal temperature sensor monitors the connector temperature. When the junction temperature exceeds 145°C, the internal control circuit and the high-side PMOS are turned off. When the junction temperature drops to 100°C, the regulator automatically restarts under the control of the soft-start circuit.

application information

The basic AOZ1014 application circuit is shown in Figure 1. Component selection is described below.

input capacitor

Input capacitors must be connected to the V pin and PGND pin of the AOZ1014 to maintain a stable input voltage and filter out pulsed input current. The voltage rating of the input capacitor must be greater than the maximum input voltage and ripple voltage.

The input ripple voltage can be approximated by the following equation:

Since the input current of a buck converter is discontinuous, the current stress on the input capacitor is another consideration when choosing capacitors. For a buck circuit, the rms value of the input capacitor current can be calculated by the following formula:

If m is equal to the conversion ratio:

The relationship between input capacitor rms current and voltage slew rate is shown in Figure 2 below. It can be seen that the current stress of C is the largest when V is half of V. The maximum current stress of C is 0.5·I.

For reliable operation and optimum performance, the input capacitor current rating must be higher than the worst-case operating conditions of Iat. Ceramic capacitors are the first choice for input capacitors due to their low ESR and high ripple current ratings. Other low ESR tantalum capacitors can also be used depending on the application circuit. When choosing ceramic capacitors, X5R or X7R type dielectric ceramic capacitors are preferred due to their better temperature and voltage characteristics. Note that capacitor manufacturers' ripple current ratings are based on a certain lifetime. Actual design requirements may require further derating.

sensor

The inductor is used to provide a constant current output when it is driven by a switching voltage. For a given input and output voltage, the inductor and switching frequency together determine the inductor ripple current, that is,

The peak inductor current is:

High inductance provides low inductor ripple current, but requires larger size inductors to avoid saturation. Low ripple current reduces inductor core losses. It also reduces the rms current through the inductor and switch, thereby reducing conduction losses. Typically, the peak-to-peak ripple current on the inductor is designed to be 20% to 30% of the output current.

When choosing an inductor, make sure it can handle peak currents without saturation even at the highest operating temperature.

The inductor accepts the highest current in the buck circuit. Conduction losses on inductors need to be checked for compliance with thermal efficiency requirements.

Coilcraft, Elytone and Murata offer surface mount sensors in different shapes and styles. The shielding inductance is small in size, and the radiated electromagnetic interference is small. But they are more expensive than unshielded inductors. The choice depends on EMI requirements, price and size.

The following table lists the inductors for some typical output voltage designs.

output capacitor

Select the output capacitor based on the DC output voltage rating, output ripple voltage specification, and ripple current rating.

The selected output capacitor must have a higher voltage rating than the maximum expected output voltage including ripple. Long-term reliability requires consideration of degradation.

The output ripple voltage specification is another important factor in selecting an output capacitor. In a buck converter circuit, the output ripple voltage is determined by the inductor value, switching frequency, output capacitor value, and ESR. It can be calculated by the following formula:

where C is the output capacitor value and ESR is the equivalent series resistance of the output capacitor.

When using a low ESR ceramic capacitor as the output capacitor, the impedance of the capacitor at the switching frequency dominates. The output ripple is mainly caused by the capacitor value and the inductor ripple current. The output ripple voltage calculation can be simplified as:

When the ESR impedance at the switching frequency dominates, the output ripple voltage is primarily determined by the capacitor ESR and inductor ripple current. The output ripple voltage calculation can be further simplified as:

To reduce the output ripple voltage over the entire operating temperature range, it is recommended to use X5R or X7R dielectric ceramic or other low ESR tantalum as the output capacitor.

In a buck converter, the output capacitor current is continuous. The rms current of the output capacitor is determined by the peak-to-peak ripple current of the inductor. The calculation method is as follows:

Usually, the ripple current rating of the output capacitor is a lesser concern due to the low current stress. When the buck inductor is chosen to be small and the inductor ripple current is large, the output capacitor will be overstressed.

loop compensation

AOZ1014 adopts peak current mode control, which is easy to use and has fast transient response. Peak current mode control eliminates the bipolar effect of the output L&C filter. This greatly simplifies the design of the compensation loop.

With peak current mode control, the buck power stage can be simplified as a one-pole-one-zero system in the frequency domain. The pole is the dominant pole and can be calculated by the following formula:

The zero is the ESR zero due to the output capacitance and its ESR. Its calculation method is as follows:

Among them, CO is the output filter capacitor; RL is the load resistance value; ESRCO is the output capacitor.

The compensation design actually obtains the desired gain and phase by changing the closed-loop transfer function of the converter. Several different types of compensation networks can be used with the AOZ1014. In most cases, a series capacitor and resistor network connected to the COMP pin sets the pole zero and is sufficient for a stable high bandwidth control loop.

In the AOZ1014, the FB pin and the COMP pin are the inverting input and output of the internal transconductance error amplifier. A series R and C compensation network connected to COMP provides one pole and one zero.

The rods are:

Among them, GEA is the transconductance of the error amplifier, which is 200•10-6 A/V; GVEA is the voltage gain of the error amplifier, 500 V/V; CC is the compensation capacitor.

The zero given by the external compensation network capacitor C and resistor R is located at:

In order to design the compensation circuit, the target crossover frequency f must be chosen as the closed loop. The system crossover frequency is where the control loop has unity gain. The crossover frequency is also known as the converter bandwidth. Generally, higher bandwidth means faster response to load transients. However, considering the stability of the system, the bandwidth should not be too high. When designing the compensation loop, the stability of the converter under all line and load conditions must be considered.

Typically, it is recommended to set the bandwidth to be less than 1/10 of the switching frequency. The AOZ1014 operates over a fixed switching frequency range of 350kHz to 600kHz. It is recommended to choose a crossover frequency less than 30kHz.

The strategy for choosing R and C is to use R to set the crossover frequency and C to set the compensator zero. Compute R with the chosen crossover frequency f:

where fC is the desired crossover frequency; VFB is 0.8V; GEA is the error amplifier transconductance, which is 200•10-6 A/V; and GCS is the current-sense circuit transconductance, which is 9.02 A/V.

Compensation capacitor C and resistor R together form zero. This zero is placed close to the dominant pole f, but below 1/5 of the chosen crossover frequency. C can be selected by:

The above equation can also be simplified to:

An easy-to-use application software that aids in designing and simulating compensation loops can be found on .

Thermal Management and Layout Considerations In the AOZ1014 buck regulator circuit, high pulse current flows through two circuit loops. The first loop starts from the input capacitor, to the VIN pin, to the LX pin, to the filter inductor, to the output capacitor and load, and back to the input capacitor through ground. When the high-side switch is turned on, current flows in the first loop. The second loop starts from the inductor, goes to the output capacitor and load, to the anode of the Schottky diode, to the cathode of the Schottky diode. When the low-side diode is turned on, current flows in the second loop.

In the PCB layout, minimizing the area of the two loops can reduce the noise of the circuit and improve the efficiency. It is strongly recommended to use a ground plane to connect the input capacitors, output capacitors and PGND pins of the AOZ1014.

In the AOZ1014 buck regulator circuit, the main power dissipating components are the AOZ1014, the Schottky diode and the output inductor. The total power consumption of the converter circuit can be measured by subtracting the output power from the input power.

Schottky's power dissipation can be approximated as:

where VFW_Schottky is the Schottky diode forward voltage drop. The power dissipation of the inductor can be calculated approximately from the output current of the inductor and the DCR.

The actual junction temperature can be calculated using the power dissipation in the AOZ1014 and the thermal impedance from junction to ambient.

The maximum junction temperature is 150°C, limiting the maximum load current capability. See the thermal rating curve for the AZZ1014's maximum load current at different ambient temperatures.

The thermal performance of the AOZ1014 is greatly affected by the PCB layout. During the design process, the user should take extra care to ensure that the integrated circuit operates under the recommended environmental conditions.

AOZ1014A is a standard SO-8 package. The AOZ1014D is a thermally enhanced DFN package that utilizes exposed thermal pads on the bottom to spread heat to the PCB metal. For optimum electrical and thermal performance, some layout tips are listed below. Figure 3 below demonstrates an example PCB layout for the AOZ1014A. Figure 4 below demonstrates an example PCB layout for the AOZ1014D.

1. Do not use thermal connections to the Vehicle Identification Number (VIN) and PGND pins. Dump the PGND pin and VIN pin with the largest copper area to help with heat dissipation.

2. The input capacitor should be connected to the VIN pin and PGND pin as much as possible.

3, the preferred ground plane. If a ground plane is not used, separate PGND from AGND and connect them at only one point to avoid PGND pin noise coupling to the AGND pin.

4. Keep the current trace from the LX pin to L to Co to PGND as short as possible.

5. Pour copper planes on all unused board areas and connect them to stable DC nodes such as VIN, GND, or VOUT.

6. The two LX pins are connected to the internal PFET drain. They are the low resistance thermal conduction paths and the noisiest switching nodes. Connect a copper plane to the LX pin to help with heat dissipation. This copper plane should not be too large or switching noise may couple to other parts of the circuit.

7. Keep sensitive signal traces away from the LX pin.

8. For DFN packages, the thermal pad must be soldered to the PCB metal. When using a multi-layer PCB, 4 to 6 thermal vias should be placed on the thermal pad and connected to the PCB metal on other layers to help dissipate heat.