Features

Synchronous Rectification ±1% Accuracy Internal Reference >90% Efficiency Input and Output Voltage Feedback 5.4V to 24V Input Voltage Range Internally Set 300kHz ±15% Oscillator 5V and 3.3V Always Output 12V Adjustable Boost Converter Boost Converter Connects to 5V Main Supply

application

Laptops and PDAs Handheld Portable Instruments

illustrate

The FAN5235 is a high-efficiency, high-precision DC/DC controller for notebook computer converters. Utilizing input and output voltage feedback control in current mode allows fast and stable loop response over a wide range of input and output voltage variations. The two mainline regulators are switched out of phase to minimize input ripple current. Current sensing efficiency based on MOSFET RDSON while allowing the use of sense resistors for high accuracy. An externally adjustable boost converter is set to generate 12V. The FAN5235 is available in a 24-pin QSOP package, as well as a 24-pin TSSOP package.

Function description

The FAN5235 is a high-efficiency, high-precision DC/DC controller for notebook and other portable applications. It provides all the voltages required by the system electronics: 5V, 3.3V, 12V, and 3.3V-always and 5V-always. Input and output voltage feedback in current mode control allows a wide range of input and output changes. The current sense is based on the MOSFET RDS, ON providing maximum efficiency while also allowing the use of high precision sense resistors. The 3.3V and 5V switching regulator outputs of the 3.3V and 5V architectures of the FAN5235 are generated from the unregulated input voltage using a synchronous buck converter. Both high-side and low-side MOSFETs are N-channel. The 3.3V and 5V switches have current sensing and use MOSFETs to set the output overcurrent threshold RDS, turn on. Each converter has a voltage-sensing feedback pin, the pin that shuts down the converter, and the boost voltage used to drive the high-side MOSFET. The following will discuss the design of the FAN5235 with reference to the diagram, showing the internal block diagram of the integrated circuit. 3.3V and 5V PWM current sensing Peak current sensing is done on the low side driver because the duty cycle is very low on the high side MOSFET. This samples the current 50ns after power-on and maintains that value for current feedback and overcurrent limiting. 3.3V and 5V PWM Loop Compensation Fan 5235's 3.3V and 5V control loop functions as follows for voltage mode and current feedback stability.

Each of them has independent voltage feedback pins, and as shown they use voltage feedforward to guarantee rejection of loop input voltage changes: ie pulse width modulation (PWM) ramp amplitude changes as a function of input voltage. The control compensation loop is done entirely internally using current mode feedback compensation. This scheme allows bandwidth and phase margin to be almost independent of output capacitance and ESR. 3.3V and 5V PWM Current Limit The 3.3V and 5V converters detect their own low-side MOSFETs respectively to decide whether to enter current limit. If the output current exceeds the current measurement limit threshold, then the converter input pulse Iout is equal to the skip mode setting limit for overcurrent (OC). After 8 clock cycles, the regulator is locked (HSD and LSD off). , it will immediately trigger the undervoltage protection, again lock the regulator off after a 2 microsecond delay (HSD and LSD off) The current limit setting resistor selection must include the tolerance of the current limit trip point, MOSFET turn-on resistance and temperature coefficient, and Ripple current, in addition to maximum output current. For example: the maximum DC output current of 5V is 5A, the MOSFET RDS, on is 17mΩ, the inductor is 5μH and the current is 5A. Due to the low RDS, on the low side the MOSFET will have a maximum temperature (ambient + self-heating) of only 75°C, at which point its RDS,on increases to 20m ohms.

The peak current is the DC output current plus the peak ripple current:

where T is the maximum period, VO is the output voltage, and L is the inductance. This current is in the low side MOSFET of 7A20mΩ=140mV. The current limit threshold is typically 150mV (135mV worst case) and R2 = 1KΩ, so this value applies. R2 may be necessary to add another 10% if additional noise margin is considered. Precision Current Limiting Precision current limiting can be achieved by placing a discrete sense resistor between the low-side power MOSFET and ground. In this case, the current limit accuracy is determined by the IC, +10%.

Shutdown (SDWN) The SDWN pin shuts down all 5 converters (+5V, +3.3V, and +12V, 5V/3.3V - always) and puts the fan 5235 in low power mode (shutdown mode). This mode of operation means using a button to switch between SDWN and VIN. Pushing the button allows (during contact) to power the 3.3V-ALWAYS with 5V - always long enough for the uC to power up to turn the SDWN pin high. Once SDWN is high, the total voltage is Enabled High if the corresponding SDN3.3 and SDN5 are on. Main 3.3V and 5V Soft Start, Sequencing and Standby Soft start of 3.3V and 5V converters is provided by an external capacitor between pin SDN3.3 (SDN5) and ground. If both SDWN and SDN3.3(SDN5) are high, if one of SDWN is off or SDN3.3(SDN5) is low. Standby mode means V power is off and V- is always on (SDWN=1 and SDN3.3=SDN5=0). Always run mode if 5V-always and 3.3V-always are required then the SDWN pin must be permanently connected to the VIN. This way, the two regulators are power as soon as they get there, and the state of the main regulator is controllable via the SDN5 and SDN3.3 pins.

3.3V Voltage Regulation The output voltage of the 3.3V converter can be up to 10% by inserting a resistor divider in the feedback. The feedback pin impedance is about 66KΩ. So, for example, to increase the output of a 3.3V converter by 10%, use a 2.21KΩ/33.2KΩ voltage divider. Note that the output of the 5V regulator cannot be adjusted. The feedback line of the 5V regulator is used internally as a 5V supply, so it cannot withstand being in series with it. 3.3V and 5V main overvoltage protection When the output voltage of the 3.3V (or 5V) converter exceeds 115% of the rated value, the converter enters the overvoltage protection mode to protect the load from damage. During operation, a dump or short circuit of the MOSFET on a heavy load can cause the output voltage to be significantly higher than the normal operating range without circuit protection. When the output exceeds the overvoltage threshold, the overvoltage comparator forces the low gate driver high and turns on the low MOSFET. This will pull down the output voltage, which will eventually fuse the battery. Once the output voltage falls below the threshold, the OVP comparator turns off. The OVP scheme also provides a soft crowbar function (bang control followed by a fuse blow) that helps with severe load transients, but does not reverse the output voltage when activated, a common problem with OVPs lock plan. The prevention of output inversion does not require a Schottky diode to pass through the load.

3.3V and 5V Undervoltage Protection When the output voltage of 3.3V or 5V falls below 75% of the nominal value, both converters enter undervoltage (UV) protection with a delay of 2usec. In undervoltage protection, the high-side and low-side MOSFETs are turned off. Once triggered, the undervoltage protection remains on until power is cycled or the SDWN pin is reset. 12V Architecture The 12V converter is a traditional non-isolated flyback (also known as a "boost" converter). The input voltage to the converter is the +5V switcher output, so +12V can only be present if +5V is present. Also, if the external MOSFET is off, the output of the +12V converter is +5V, not zero. In turn will provide a non-zero output for the 12V regulator. To completely turn off the 12V regulator, an external P-channel MOSFET or LDO regulator with switching control can be used. If the LDO uses 12V, the boost should use an external resistor to set the converter to 13.2V to split the network. The 12V loop compensated 12V converter should operate in discontinuous conduction mode.

In this mode, if a capacitor with an appropriate ESR value is selected. 68 cubic feet of tantalum with a ripple current rating of 500mA, 95mΩ is recommended here. 12V Protection The 12V converter is protected against overvoltage. If the 12V feedback is 10–15% higher than the nominal value, the comparator forces the MOSFET off until the voltage falls below the comparator threshold. The 12V converter is also protected against overcurrent. If the short circuit pulls the output below 9V, all switching converters go into UV protection after a 2 microsecond delay. UV protection, all mosfets are off. Once the UV protection is triggered, it remains on until the input power is recycled or the SDWN is reset. 12V Soft Start and Sequencing The 12V output starts simultaneously with the 5V output. A softly rising 5V output automatically produces a softly rising 12V output. The duty cycle of the 12V PWM is limited to prevent excessive current consumption. The voltage of the 12V supply must be higher than the UVLO limit (9V) (3.75V) when 5V is above its UVLO to avoid soft start.

5V/3.3V - ALWAYS RUNNING 5V - ALWAYS powered by onchip linear regulators or via the VFB pin of a 5V switching power supply. When the 5V switching power supply is turned off, or its output voltage is not within tolerance, the 5V-ALWAYS switch is turned on and the linear regulator is turned on. When the 5V switching power supply is running and the output voltage is within the specified range, the linear regulator is turned off, and the switch is turned on. The switch has a sufficiently low resistance 5V at maximum current draw - always powered and the output voltage regulated within spec. 3.3V - always generated by the linear regulator internally connected to 5V - always. The purpose of the two total supplies (the combined current is specified not to exceed 50 mA) is to provide power to the system microcontroller (Class 8051) and other integrated circuits that require a backup power supply. Microcontroller also because the other IC can operate on 5V or 3.3V all the time, so the FAN5235 provides both. 5V/3.3V-Always protected Both internal linear regulators are current-limited under-voltage protection. Once all trigger protection outputs are off until power cycle or SDWN shutdown reset. Power is good when both PWM Buck converters are above the specified threshold. No other regulator is monitored by force very well. At least 10 microseconds (Tw) when feeling down. See Fig.

The error amplifier output voltage is clamped during load transients, allowing the error amplifier voltage to advance at full speed. After two clock cycles, if the amplifier is still out of voltage range, the duty cycle (DC) is clamped. The DC clamp automatically limits overcurrent under abnormal conditions, including short circuits:

Thermal Shutdown If the die temperature of the fan 5235 exceeds a safe limit, the IC shuts down automatically. When the over temperature (OT) event ends, the IC resumes normal operation. There is a 25°C thermal hysteresis between shutdown and startup. Input UVLO If the input voltage is below the UVLO threshold, the FAN5235 automatically shuts down as long as the input voltage is below the threshold.

The MOSFET selection has designed the notebook application circuit shown in the figure to operate from an input voltage of 5.4-24V. This wide input range helps determine the mosfet for 3.3V and 5V converters because the high side is when the MOSFET is on (Vout/Vin) and the low side is when MOSFET 1 – (Vout/Vin). The maximum and minimum values are shown in Table 2:

The maximum duty cycle of all four MOSFETs is greater than 50%. Therefore, it is necessary to approximate all four sizes to be the same. The maximum current mode of the 3.3V and 5V Schottky selection converters operating in the PFM determines the Schottky selection. As shown in the application, because at currents up to 28mV*(17.5KΩ/10KΩ)/35mΩ=1.4A, the diode (with 24V input) will perform 86% of the cycle (from Table 2). So its average current is 1.4A*0.86=1.2A, requiring a Schottky current rating greater than 1A. See Table 1 for 3.3V and 5V inductor selections. See Table 1 for 3.3V and 5V output cap selections. 12V Component Selection Inductor, diode, and output capacitor calculations are complex for +12V output feedback and depend on output power as well as efficiency. Excel See Application Bulletin AB-19 Spreadsheet Calculation Tool. See also Table 1. Input Capacitor Selection The choice of input capacitor depends on the ripple current rating. With two converters operating in parallel at different loads, the calculation of the input ripple current is complex; see Application Bulletin AB-19 Calculation Tool for Excel Spreadsheets.