Designed for bidirectional pulse width modulation (PWM) current control of inductive loads, the A353S-capable of continuous output current ±1.3 A and operating voltage of 50 V. An internal fixed off-time PWM current control circuit can be used to regulate the maximum load current to the desired value. The peak load current limit is set by user selection of the input reference voltage and external sense resistor. The fixed off-time pulse duration is set by a user-selected external RC timing network. Internal circuit protection includes hysteretic thermal shutdown, TVS diode, and cross-current protection. No special power-up sequence is required.

With the enable input held low, the phase input controls the load current polarity by selecting the appropriate source and sink driver pair. The mode input determines whether the pwm current control circuit operates in slow current decay mode (source driver switching only selected) or fast current decay mode (source and sink switching selected). A user-selectable blanking window prevents false triggering of the PWM current control circuit. When the enable input is held high, all output drivers are disabled. Sleep mode is provided to reduce power consumption.

The brake function is enabled when logic low is applied to the brake input. This override enables and phase to turn off both source drivers and turn on both sink drivers. The braking function can be used to dynamically brake brushed DC motors.

A3953S - Two power packs are available: a 16-pin dual in-line plastic pack with a copper heat sink, and a 16-pin plastic SOIC with a copper heat sink. For both package types, the power label is at ground potential and electrical isolation is not required.

feature

±1.3 A continuous output current Sleep (low current 50 V output rated voltage consumption) mode 3 V to 5.5 V logic supply voltage Internal transients Internal PWM current control Suppression diode saturation sink driver (below 1A) Internal thermal fast and slow current Decay mode shuts down the circuit; cross-current and UV protection for automotive use.

Function description

Internal PWM current control during forward and reverse operation. A3953S - Contains a fixed untimed pulse width modulation (PWM) current control circuit that can be used to limit the load current to the desired value. The peak value of the current limit (ITRIP) is set by the selection of an external current sense resistor (RS) and the reference input voltage (VREF). An internal circuit compares the voltage on the external sense resistor to the voltage on the reference input (REF), resulting in an approximate transconductance function of:

where iso is the offset due to the base drive current.

In forward or reverse mode, the current control circuit limits the load current as follows: When the load current reaches itrip, the comparator resets a latch that turns off the selected source driver or the selected sink and source driver pairs, depending on whether the device operates in slow or fast current decay mode, respectively.

In slow current decay mode, the selected source driver is disabled; the load inductance causes current to recirculate through the sink driver and ground clamp diode. In fast current decay mode, the selected sink and source driver pairs are disabled; the load inductor causes current to flow from ground to the load supply through the ground clamp and flyback diode.

The user selects an external resistor (RT) and capacitor (CT) to determine the period of time the driver remains disabled (TOFF = RT x CT) (see "RC Fixed Off Time" below). At the end of the rc interval, the driver is enabled, allowing the load current to increase again. The pulse width modulation cycle repeats, maintaining the peak load current at the desired value (see Figure 2).

Brake Operation - Mode Input High. The braking circuit turns off the two source drivers and turns on the two sink drivers. For DC motor applications, this causes a short circuit to the motor's back-EMF voltage, resulting in a dynamic brake motor current flow. If the back EMF voltage is large and there is no pwm current limit, the load current can be increased to a value close to the stalled state. To limit the current, the pwm circuit will disable the conducting receiver driver when the itrip level is reached. The energy stored in the motor inductance is released into the load power supply, causing the motor current to decay.

In the case of forward/reverse operation, the driver is enabled after a time given by toff=rt x ct (see "rc fixed off time" below). Depending on the back EMF voltage (proportional to the deceleration of the motor), the load current may increase again to ITRIP. If so, the PWM cycle is repeated to limit the peak load current to the desired value.

During braking, when the mode input is high, the peak current limit can be approximated as:

NOTE: Because the kinetic energy and load inertia stored in the motor are converted into current to charge the bulk capacitance of the VBB power supply (power output and decoupling capacitors), care must be taken to ensure that the capacitors are sufficient to absorb the energy without exceeding the amount of energy connected to the motor power supply rated voltage of any device.

Brake Operation - Mode Input Low. During braking, when the mode input is low, the internal current control circuit is disabled. Therefore, care should be taken to ensure that the current of the motor does not exceed the rating of the device. Braking current can be measured with an oscilloscope using a current probe connected to one of the motor leads, or if the motor's back EMF voltage is known, by:

RC fixed closing time. An internal PWM current control circuit uses one trigger to control how long the driver remains off. The one-shot time toff (fixed off-time) is determined by selecting an external resistor (rt) and capacitor (ct) connected in parallel from the rc timing terminal to ground. The fixed cutoff time, in the range of CT = 470 pF to 1500 pF and rT = 12 K to 100 K, is approximately:

The circuit operates as follows: When the PWM latch is reset by the current comparator, the voltage on the RC terminal will begin to decay from approximately 0.60VCC. When the voltage on the RC terminal reaches about 100 volts, the PWM latch is set, thus enabling the driver(s).

Reinforced concrete cutting. In addition to determining the fixed off time of the pwm control circuit, the CT component also sets the comparator blanking time. This function will blank the output of the comparator when the internal current control circuit (or the phase, brake or enable input) switches the output. The comparator output is shielded to prevent false overcurrent detection due to reverse recovery current of the clamp diode and/or switching transients related to distributed capacitance in the load.

During internal PWM operation, at the end of the TOFF time, the output of the comparator is blanked and CT begins to be charged from about 0.22VCC by the internal current source of about 1mA. The comparator output remains blank until the voltage on CT reaches approximately 0.60VCC.

When the phase input transitions, CT is discharged to near ground during the crossover delay time (which exists to prevent the source and sink drivers from turning on at the same time). After the crossover delay, CT is charged by an internal current source of approximately 1 mA. The comparator output remains blank until the voltage on CT reaches approximately 0.60VCC.

When the device is disabled, the CT is discharged close to ground via the enable input. When the device is restarted, the CT is charged by an internal current source of approximately 1 mA. The comparator output remains blank until the voltage on CT reaches approximately 0.60 cm3.

For 3.3V operation, the minimum recommended value for CT is 680pF5%. For 5.0V operation, the minimum recommended value for CT is 470 pF 5%. ±± These values ensure that the blanking time is sufficient to avoid false tripping of the comparator under normal operating conditions. For optimal regulation of the load current, the CT value above is recommended, and the RT value can be adjusted to determine TOFF.

Load current regulation using internal pwm current control circuit

When the device is operating in slow current decay mode, there is a minimal limitation that the pwm current control circuit can regulate the load current. The limit is the minimum duty cycle, which is a function of the user-selected toff value and the minimum on-time pulse ton(min)max that occurs each time the pwm latch is reset. If the motor is not spinning (such as when a stepper motor is in hold/brake mode, when stalling, or brushing a DC motor at startup), the worst-case value for current regulation can be approximated as:

When the motor rotates, the generated back EMF affects the above relationship. For brushed DC motor applications, current regulation is improved. For stepper motor applications, the effect is more complex when the motor rotates. The following chapter on stepper motors will discuss this issue.

The following procedure can be used to evaluate the worst-case slow current decay internal PWM load current regulation in a system:

Set VREF to 0 volts. With the load connected and the PWM current control operating in slow current decay mode, use an oscilloscope to measure the low output (pick-up) time of the chopped output. This is the typical minimum turn-on time for the device (ton (minute) type). CT should then be increased until the ton (min) measurement is equal to the ton (min) max specified in the electrical characteristics table. When the new value of CT is set, the value of RT should be decreased so that the value of TOFF=RT x CT (artificially increased value of CT) is equal to the nominal design value. Worst-case load current regulation can be measured under system operating conditions.

When the device is enabled, the internal current control circuit will activate and can be used to limit the load current in slow current decay mode.

For applications that pwm the enable input and require the internal current limiting circuit to operate in fast decay mode, the enable input signal should be inverted and connected to the mode input. This prevents the device from switching to sleep mode when the enable input is low.

Phase modulation. Switching the phase termination selects which sink/source pair is enabled, resulting in a duty cycle-variant and always continuous load current. This has an added benefit in bidirectional brushed DC servo motor applications, as the transfer function between the duty cycle of the phase input and the average voltage applied to the motor is smaller than that of enabling pwm control (which produces discontinuous current at low current levels) case is more linear. See "DC Motor Applications" below for more information.

Synchronous fixed frequency pulse width modulation. Using the simple circuit shown in Figure 3, the internal PWM current control circuits of multiple A3953S devices can be synchronized. The 555 IC can be used to generate the reset pulse/blanking signal (t1) and pwm period (t2) of the device. The value of T1 should be at least 1.5 ms. When used in this configuration, the RT and CT components should be omitted. The phase and enable inputs should not take the pwm of this circuit configuration as there is no blanking synchronous to their transitions.

other information. A logic high applied to the enable and mode terminals puts the device into sleep mode to minimize current consumption when not in use.

Internally generated dead time prevents cross currents when switching phases or braking.

Thermal protection circuitry will shut down all drivers if the connector temperature reaches 165°C (typical). This is done only to prevent the device from failing due to excessive junction temperature, it does not imply that the output is allowed to short circuit. The thermal shutdown circuit has a hysteresis of about 15°C.

Application Notes

current sensing. Due to the drive turn-off delay, the actual peak load current (IPEAK) will be higher than the calculated value for ITRIP. The overshoot can be approximated as:

Where vbb is the power supply voltage of the motor, vbemf is the load back EMF voltage, rload and lload are the load resistance and inductance respectively, and tpwm (off) is specified in the electrical characteristics table.

The reference terminal has a maximum input bias current of ±5µA. This current should be considered when determining the impedance of the external circuit that sets the reference voltage value.

To minimize current sensing inaccuracies caused by the ground trace ixr dropping, the current sensing resistor should have a separate return path to the device ground terminal. For low value sense resistors, the I x R drop in the printed wiring board can be large and should be considered. Receptacles should be avoided because the contact resistance of the receptacle will cause a change in the rms value of RS.

In general, the larger the value of rs, the smaller the above effects, but will result in overheating of the sense resistor and excessive power loss. The value of RS chosen should not result in an absolute maximum voltage rating of 1 V (0.4 V for VCC = 3.3 V operation), for sensory terminations that would exceed.

The current sense comparator functions all the way to ground, allowing the device to be used for microstepping, sinusoidal and other different current profile applications.

thermal factor. For reliable operation, it is recommended to keep the maximum junction temperature below 110°C to 125°C. Junction temperature is best measured by connecting a thermocouple to the device's power supply Tab/ButWin and measuring the TAB temperature, TTAB. The junction temperature can then be calculated approximately using the formula:

where VF can be selected from the electrical specification table for a given ILOAD level. The value of R is given in the package thermal resistance table for the corresponding package.

By connecting a length of printed circuit board copper (usually 6 to 18 square centimeters) to the device's batwing terminals, the power consumption of the batwing package can be increased by 20 to 30 percent.

In applications operating at high load currents and/or high duty cycles, thermal performance can be improved by adding an external diode in parallel with the internal diode. In internal pwm slow decay applications, only two ground clamp diodes need to be added. For internal fast decay PWM, or external phase or enable input PWM applications, all four external diodes should increase the maximum junction temperature reduction.

printed circuit board layout. The load power terminal VBB should be separated from the electrolytic capacitor (more than 47μF is recommended), as close as possible to the device. To minimize the effect of the system ground ixr drop on the logic and reference input signals, the system ground should have low resistance back to the motor supply voltage.

Fixed closing time selection. As the toff value increases, the switching loss decreases, the low-level load current regulation improves, the emi decreases, the pwm frequency decreases, and the ripple current increases. The toff value can be chosen to optimize these parameters. For applications involving audible noise, typical values of toff are chosen in the range of 15 ms to 35 ms.

Stepper motor applications. This mode termination can be used to optimize device performance in microstepping/sine stepper motor drive applications. When the load current increases, a slow decay method is adopted to limit the switching loss of the device and the iron loss of the motor. This also increases the maximum rate at which the load current can increase (compared to fast decay) because the decay rate is slow during TOFF. When the load current decreases, use the fast decay mode to regulate the load current to the desired level. This prevents the back EMF voltage of the stepper motor from causing tailing of the current waveform.

In stepper motor applications where a constant current is applied to the load, slow decay mode pwm is often used to limit switching losses in the device and iron losses in the motor.

DC motor applications. In a closed loop system, the speed of the DC motor can be controlled by the pwm of the phase or enable input or by changing the reference input voltage (ref). In digital systems (microprocessor controlled), the pwm of the phase or enable input is often used to avoid a variable analog voltage reference. In this case, the DC voltage at the REF input is usually used to limit the maximum load current.

In DC servo applications requiring precise positioning at low or zero speed, the pwm of the phase input is usually chosen. This simplifies the servo control loop because the transfer function between the phase input duty cycle and the average voltage applied to the motor is more linear than it would be if pwm control was enabled (which produces discontinuous current at low current levels).

Using bidirectional DC servo motor, the phase terminal can be used for mechanical direction control. Similar to a dynamic brake motor, a sudden change in the direction of a rotating motor produces a current generated by the back EMF. The resulting current will depend on the mode of operation. The maximum load current can be generated if the internal current control circuit is not used.

iload=(vbemf+vbb)/rload, where vbemf is proportional to the motor speed. If the internal slow current decay control circuit is used, the resulting maximum load current can be approximated by ILoAD = VBEMF/RLADE. In both cases, care must be taken to ensure that the maximum ratings of the device are not exceeded. If the internal fast current decay control circuit is used, the load current will be regulated to the following given value:

Note: In fast current decay mode, when the direction of the motor is suddenly changed, the kinetic energy and load inertia stored in the motor are converted into current that charges the VBB power supply capacitors (power output and decoupling capacitors). Care must be taken to ensure that the capacitance is sufficient to absorb energy without exceeding the voltage rating of any equipment connected to the motor power supply.