Features

Complete PWM Power Control Circuitry Uncommitted Outputs for 200 mA of Sink or Source Current Output Control Selection of Single-Ended or Push-Pull Operation The device provides a stable 5V reference voltage with a 5% tolerance circuit structure allows easy synchronization of applications

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3 Description: The TL494 device contains all the functions needed to build a pulse width modulation (PWM) control circuit on a single chip. The device is primarily designed for power control, with the flexibility to customize the power control circuit for specific applications.

The TL494 device includes two error amplifiers, an on-chip adjustable oscillator, a dead-time control (DTC) comparator, a pulse steering control flip-flop, a 5-volt, 5% precision regulator, and output control circuitry.
The common-mode voltage range of the error amplifier is -0.3V to VCC-2V. Dead-time control comparators with fixed compensation provide approximately 5% dead time. By terminating RT to the reference output and providing a sawtooth input to CT, the on-chip oscillator can be bypassed or common circuits in synchronous multi-rail supplies can be driven.
Uncommitted output transistors provide common emitter or emitter follower output capability. The TL494 device offers push-pull or single-ended output operation, selectable through the output control function. The structure of the device prohibits two pulse outputs during push-pull operation.
The TL494c device is characterized for operation from 0°C to 70°C. The TL494i device is characterized for operation in temperatures ranging from -40°C to 85°C.

Overview
The design of the TL494 not only contains the main building blocks required to control a switching power supply, it also solves many fundamental problems and reduces the number of additional circuits required in the overall design. TL494 is a fixed frequency pulse width modulation (PWM) control circuit. The modulation of the output pulse is done by comparing the sawtooth wave generated by the internal oscillator on the timing capacitor (CT) with either of the two control signals. When the sawtooth voltage is greater than the voltage control signal, the output stage is enabled. As the control signal increases, the time of the sawtooth input decreases and therefore the output pulse duration decreases. The pulse steering flip-flop alternately directs the modulated pulse to each of the two output transistors.

Function description
1 5-V reference regulator
The 5-V reference regulator output inside the TL494 is the reference pin. In addition to providing a stable reference, it acts as a pre-regulator and establishes a stable power supply from which to power the output control logic, pulse steering flip-flop, oscillator, dead time control comparator, and PWM comparator. The regulator employs a bandgap circuit as its primary reference to maintain thermal stability of less than 100 mV over the operating free air temperature range of 0°C to 70°C. Short-circuit protection is provided to protect the internal reference and preregulator; a load current of 10 mA is available for additional bias circuits. The reference is internally programmed to have an initial accuracy of ±5% and is stable to less than 25 mV of change over an input voltage range of 7 V to 40 V. For input voltages less than 7 V, the regulator saturates and tracks within 1 V of the input voltage.

Functional Description (continued)
Oscillator The oscillator provides a positive sawtooth wave to dead-time and PWM comparators for comparison with various control signals. The frequency of the oscillator is programmed by selecting timing elements RT and CT. The oscillator charges the external timing capacitor CT with a constant current value determined by the external timing resistor RT. This produces a linear ramp voltage waveform. When the voltage of the current transformer reaches 3 volts, the oscillator circuit discharges it and restarts the charge cycle. The charging current is determined by the following formula:
However, the oscillator frequency is only equal to the output frequency for single-ended applications. For push-pull applications, the output frequency is half the oscillator frequency.
Single-ended applications:
Push-pull application:
Dead Time Control The dead time control input provides control of the minimum dead time (off time). When the input voltage is greater than the oscillator's ramp voltage, the output of the comparator inhibits switching transistors Q1 and Q2. An internal offset of 110 mV ensures a minimum dead-time of ~3% when the dead-band control input is grounded. Additional dead time can be applied by applying voltage to the dead time control input. This provides linear control of dead time from a minimum of 3% to 100% when the input voltage is varied from 0 V to 3.3 V, respectively. With full range control, the output can be controlled from an external source without disturbing the error amplifier. The dead time control input is a relatively high impedance input (II < 10µA) and should be used when additional control of the output duty cycle is required. However, for proper control, the input must be terminated. An open circuit is an undefined condition.
Comparator The comparator is offset from the 5-V reference regulator. This provides isolation from the input power supply for improved stability. The input to the comparator does not exhibit hysteresis, so protection against false triggering must be provided near the threshold. The comparator has a 400 ns response time from either control signal input to the output transistor, with only 100 mV of overdrive. This ensures positive control of the output for half a cycle when operating within the recommended 300 kHz range.

Functional Description (continued)
Pulse Width Modulation (PWM)
The comparator also provides modulation control of the output pulse width. To do this, the ramp voltage of the timing capacitor CT is compared with the control signal at the output of the error amplifier. The timing capacitor input contains a series diode omitted from the control signal input. This requires the control signal (error amplifier output) to be ~0.7V larger than the voltage on CT to suppress the output logic and ensure maximum duty cycle operation without requiring the control voltage to drop to true ground potential. The output pulse width varies from 97% of the period to 0, as the voltage at the error amplifier output varies from 0.5 V to 3.5 V, respectively.
Error Amplifiers - Both high gain error amplifiers receive their offset from the vi supply rail. This allows a common-mode input voltage range of -0.3 V to 2 V, which is less than vi. Both amplifiers are characterized as single-ended, single-supply amplifiers, as each output is active high only. This allows each amplifier to be pulled up independently to reduce the need for output pulse width. When the two outputs are simultaneously ORed at the inverting input node of the PWM comparator, the amplifier requiring the smallest pulse output dominates. The amplifier output is biased low by the current sink to provide maximum pulse width output when both amplifiers are biased.
Output Control Input The output control input determines whether the output transistors work in parallel or push-pull. This input is the power supply for the pulse steering trigger. The output control input is asynchronous and controls the output directly, independent of the oscillator or pulse steering trigger. Input conditions are fixed conditions defined by the application. For parallel operation, the output control input must be grounded. This disables the pulse steering trigger and disables its output. In this mode, the pulses seen at the output of the dead-time control/PWM comparator are transmitted in parallel by two output transistors. For push-pull operation, the output control input must be connected to the internal 5V reference voltage regulator. In this case, each output transistor is alternately enabled by a pulse steering flip-flop.
output transistor
There are two output transistors on the TL494. Both transistors are configured as open collector/open emitter, and each transistor can sink or source up to 200mA. In the common emitter configuration, the saturation voltage of the transistor is less than 1.3 V, while in the emitter follower configuration, the saturation voltage of the transistor is less than 2.5 V. The outputs are protected against excessive power dissipation to prevent damage, but sufficient current limit is not used to allow them to operate as current source outputs.
Device Functional Mode The TL494 operates in single-ended or parallel mode when the output control pin is grounded. When the output control pin is connected to VREF, the TL494 operates in normal push-pull operation.

APPLICATIONS AND IMPLEMENTATION NOTE: The information in the following application sections is not part of the TI component specification, and TI does not warrant its accuracy or completeness. TI's customers are responsible for determining the suitability of components for their purpose. Customers should verify and test their design implementation to confirm system functionality.
Application Information The following design example uses the TL494 to create a 5-V/10-A power supply. This application is excerpted from application SLVA001.
typical application

Typical Applications (continued)
Design requirements 8226 ;vi=32 V•vo=5 V•io=10 A•fosc=20 kHz Switching frequency•vr=20 mV peak-to-peak (vripple)•Δil=1.5-A Inductor current variation Detailed design procedure Input power supply This The power supply's 32-V DC power source uses a 120-V input 24-V output transformer rated at 75 VA. A 24-V secondary winding powers a full-wave bridge rectifier, followed by a current-limiting resistor (0.3Ω) and two filter capacitors

Typical Applications (continued)
Error Amplifier Section
The 5V reference voltage inside the TL494 is divided into 2.5V by r3 and r4. The output voltage error signal is also divided into 2.5 V by r8 and r9. If the output must be precisely adjusted to 5.0 V, a 10-kΩ potentiometer can be used in place of R8 to adjust.
To improve the stability of the error amplifier circuit, the output of the error amplifier is fed back to the inverting input through RT, reducing the gain to 101.
The current limiting amplifier supply is designed for a 10-A load current and 1.5 A IL swing, so the short circuit current should be:
Resistors R1 and R2 set a reference voltage of about 1 V on the inverting input of the current-limiting amplifier. Resistor R13 is in series with the load and applies 1 V to the non-inverting terminal of the current limiting amplifier when the load current reaches 10 A. The output pulse width is reduced accordingly. Equation 11 calculates the value of R13.
Soft-Start and Dead Time To reduce the stress on the switching transistor during startup, it is necessary to reduce the startup surge that occurs when the output filter capacitors are charged. Availability of dead-band control makes soft-start circuit implementation relatively simple

The soft-start circuit allows the pulse width at the output to slowly increase by applying a negative slope waveform to the dead-time control input (pin 4). Initially, capacitor C2 forces the dead-time control input to follow the 5-V regulator, thereby disabling the output (100% dead time). As the capacitor is charged through R6, the output pulse width slowly increases until the control loop accepts the command. With a resistor ratio of R6 and R7 of 1:10, the voltage at pin 4 after startup is 0.1×5 V or 0.5 V.
Soft-start times are typically in the range of 25 to 100 clock cycles. If 50 clock cycles at 20 kHz switching frequency are selected, the soft-start time is:
Then, the value of the capacitor is determined by the following formula:
This helps eliminate any erroneous signals that the control circuit might generate when power is applied.

Output Capacitor Calculation After calculating the filter inductance, calculate the output filter capacitor value to meet the output ripple requirements. An electrolytic capacitor can be modeled as a series connection of inductance, resistance, and capacitance. To provide good filtering, the ripple frequency must be well below the frequency where series inductance becomes important. Therefore, the two parts of interest are capacitance and effective series resistance (ESR). The maximum ESR is calculated from the specified relationship between peak-to-peak ripple voltage and peak-to-peak ripple current.
The minimum capacitance of C3 required to maintain the VO ripple voltage below the 100 mV design target is calculated from Equation 15:
The 220 mf, 60-v capacitor was chosen because it has a maximum ESR of 0.074Ω and a maximum ripple current of 2.8 A.
Transistor Power Switch Calculation Transistor power switch consists of NTE153 PNP drive transistor and NTE331 NPN output transistor. The two power supplies are connected in a PNP Hybrid Darlington circuit configuration

Hybrid Darlington circuits must saturate at IO+ΔIL/2 or a maximum output current of 10.8 A. The 10.8 A Darlington HFE must be high enough not to exceed the TL494's 250 mA maximum output collector current. According to the published NTE153 and NTE331 specifications, the minimum driving force required for the power switch is calculated from Equation 16, and Equation 18 is 144 mA:
The R10 value is calculated as follows:
Based on these calculations, choose the closest standard resistor value of 220Ω for R10. Resistors R11 and R12 allow the charge carriers in the switching transistor to discharge when turned off.
the electricity

Power Recommendations
The TL494 is designed to operate over an input voltage range of 7 V to 40 V. This input supply should be well regulated. If the input supply is more than a few inches away from the device, additional bulk capacitors may be required in addition to ceramic bypass capacitors. A tantalum capacitor with a value of 47µF is a typical choice, but this may vary depending on the output power.
Layout Guidelines Always try to use ferrite type closed core low EMI inductors. Some examples are toroidal and core-spun inductors. Open cores can be used if they have lower EMI characteristics and are a little further away from low power traces and components. If using an open core, also keep the electrodes perpendicular to the PCB. Rod cores usually make the most unwanted noise.
Feedback Tracking Try to keep the feedback track as far away from the inductor and noise power tracks as possible. You also want the feedback tracking to be as direct as possible and a little thick. These two issues sometimes involve a trade-off, but keeping it away from inductive EMI and other noise sources are the more critical issues. Run a feedback trace on the PCB opposite the inductor, separating the two inductors by a ground plane.
Input/Output Capacitors When using low value ceramic input filter capacitors, it should be as close as possible to the VCC pin of the IC. This will remove as many trace inductance effects as possible and give a cleaner voltage supply to the internal IC rails. Some designs also require the use of a feedforward capacitor from the output to the feedback pin, usually for stability reasons. In this case it should also be as close to the IC as possible. Using surface mount capacitors also reduces lead length and reduces the likelihood of noise coupling into the effective antenna created by through-hole components.
Compensation Components External compensation components for stabilization should also be placed near the IC. Surface mount components are recommended here for the same reasons discussed for filter capacitors. These also shouldn't be very close to the sensor.
Traces and ground planes keep all power (high current) traces as short, straight and thick as possible. On a standard PCB, it is best practice to have an absolute minimum of 15 mils (0.381 mm) per amp for the traces. The inductor, output capacitor, and output diode should be as close as possible. This helps reduce the electromagnetic interference radiated by the power tracking due to the high switching currents passing through them. This will also reduce lead inductance and resistance, thereby reducing noise peaking, ringing, and resistive losses that create voltage errors. The grounds of the integrated circuit, input capacitors, output capacitors, and output diodes (if applicable) should be connected directly to the ground plane. It's also a good idea to have a ground plane on both sides of the PCB. This will reduce noise by reducing ground loop errors and absorbing more EMI radiated by the inductor. For multilayer boards with more than two layers, a ground plane can be used to separate the power plane (where the power traces and components are) and the signal plane (where the feedback, compensation, and components are) to improve performance. On multilayer boards, vias are needed to connect the tracks and the different planes. If the trace needs to conduct a lot of current from one plane to another, it is best to use a standard via per 200 mA of current. Arrange the components so that the switching current loops bend in the same direction. Because of the way switching regulators work, there are two power states. One state is when the switch is open, and one state is when the switch is closed. In each state, there is a current loop made up of the power components that are currently conducting. Position the power components so that the current loop conducts in the same direction in each of the two states. This prevents track-induced magnetic field reversal between two half-cycles and reduces radiated EMI.

The source demonstrates the flexibility of the TL494 PWM control circuit. This power supply design illustrates the many power control methods offered by the TL494, as well as the versatility of the control circuit.

layout example