3.5V to 28V Input Voltage Range

The output voltage is adjustable down to 0.8V . The TPS54231 is a 28V, 2A non-synchronous step-down converter with integrated low RDS(on) high-side 8226 ; integrated 80 mΩ high-side MOSFET supports the MOSFET. To improve efficiency at light loads, continuous output current pulses of up to 2A™ automatically skip the Eco-Mode function. At light loads, the pulses are activated for high efficiency. Additionally, the 1µA Shutdown Power™ skips Eco-mode current allowing the device to be powered using a fixed 570kHz switching frequency in battery applications. Current-mode control Internal slope compensation simplifies external typical 1µA shutdown quiescent current compensation calculations and reduces components to allow the use of ceramic outputs. Adjustable slow-start limiting inrush current counts with programmable UVLO threshold capacitors. A resistive divider programs the hysteresis input undervoltage lockout. Overvoltage Overvoltage Transient Protection Transient protection circuits limit voltage overshoot cycle-by-cycle current limit, start-up and frequency folding during transient conditions. Iterative and thermal shutdown protection Cyclic current limiting schemes, frequency folding and consumer applications such as set-top boxes, CPE equipment, LCD displays, peripherals and battery chargers Industrial and automotive audio power supplies 5V, 12V and 24V distributed power systems

Functional block diagram

Typical Characteristics (continued) Overview

The TPS54231 is a 28V, 2A, step-down (buck) converter with integrated high-side n-channel MOSFET. To improve performance during line and load transients, the device implements constant frequency, current-mode control, reduces output capacitance, and simplifies external frequency compensation design. The preset switching frequency of the TPS54231 is 570 kHz.

The TPS54231 requires a minimum input voltage of 3.5V for proper operation. The EN pin has an internal pull-up current source that can be used to adjust the input voltage under-voltage lockout (UVLO) with two external resistors. Additionally, when the EN pin is floating, the pull-up current provides default conditions for device operation. Operating current is typically 75µA, with no switching and no load. When the device is disabled, the supply current is typically 1 µA.

Integrated 80 mΩ high-side MOSFETs allow high-efficiency power supply designs with continuous output currents up to 2 A.

The TPS54231 reduces external component count by integrating a bootstrap charge diode. The bias voltage for the integrated high-side MOSFET is provided by an external capacitor on the PH pin. The bootstrap capacitor voltage is monitored by a UVLO circuit, which turns off the high-side MOSFET when the voltage falls below a preset threshold of 2.1v. The output voltage can be stepped down as low as the reference voltage.

By adding external capacitors, the slow-start time of the TPS54231 can be adjusted, enabling flexible output filter selection.

To improve efficiency under light load conditions, the TPS54231 typically enters a special pulse-skipping Eco modeTM when the peak inductor current falls below 100ma.

Frequency folding reduces the switching frequency during startup and overcurrent conditions, helping to control the inductor current. Thermal shutdown provides additional protection in fault conditions.

detailed description

Fixed frequency PWM control

The TPS54231 uses fixed frequency, peak current mode control. The internal switching frequency of the TPS54231 is fixed at 570kHz.

ecological model

The TPS54231 is designed to operate in Pulse Skip Eco ModeTM at light load currents to improve light load efficiency. When the peak inductor current is typically below 100mA, the COMP pin voltage typically drops to 0.5V and the device enters Eco modeTM. When the device is in Eco modeTM, the COMP pin voltage is internally clamped to 0.5V, preventing high-side integrated MOSFET switching. In order for the COMP pin voltage to rise above 0.5v and exit Eco modeTM, the inductor current peak must rise above 100ma. Since the integrated current comparator only captures the peak inductor current, the average load current entering Eco ModeTM varies with the application and external output filter.

Voltage Reference (Vref)

The voltage reference system stabilizes the output of the bandgap circuit by regulating temperature, resulting in an initial accuracy voltage reference of ±2% (±3.5% over temperature). A typical reference voltage design is 0.8V.

Boot voltage (boot)

The TPS54231 has an integrated boot regulator and requires a 0.1µF ceramic capacitor between the boot and PH pins to provide the gate drive voltage for the high-side MOSFET. Ceramic capacitors with X7R or X5R grade dielectrics are recommended because of their stable characteristics over temperature and voltage. To increase the voltage drop, the TPS54231 is designed to operate at 100% duty cycle as long as the power-on to PH pin voltage is typically greater than 2.1V.

Enable and Adjustable Input Undervoltage Lockout (VIN UVLO)

The EN pin has an internal pull-up current source that provides the default conditions for TPS54231 operation when the EN pin is floating.

The TPS54231 is disabled when the VIN pin voltage is below the internal VIN UVLO threshold. It is recommended to use an external VIN UVLO to add hysteresis unless VIN is greater than (VOUT+2V). To adjust the VIN UVLO with hysteresis, use an external circuit connected to the EN pin as shown in Figure 12. Once the EN-pin voltage exceeds 1.25v, add an additional 3µA of hysteresis. Use Equation 1 and Equation 2 to calculate the required resistor value for the desired VIN UVLO threshold voltage. VSTART is the input start threshold voltage, VSTOP is the input stop threshold voltage, and VEN is the enable threshold voltage of 1.25 V. VSTOP should always be greater than 3.5 V.

Error amplifier

The TPS54231 has a transconductance amplifier for the error amplifier. The error amplifier compares the VSENSE voltage to an internal effective voltage reference provided at the input of the error amplifier. In normal operation, the transconductance of the error amplifier is 92μA/V. The frequency compensation component is connected between the compressor pins and ground.

slope compensation

To prevent subharmonic oscillations when operating the device at duty cycles greater than 50%, the TPS54231 adds built-in slope compensation, which is a compensated ramp for the switch current signal.

Current Mode Compensation Design

To simplify the design effort using the TPS54231, Table 1 lists typical designs for common applications. For designs using ceramic output capacitors, it is recommended to appropriately reduce the ceramic output capacitors when performing stability analysis. This is because as the applied voltage increases, the actual ceramic capacitance drops significantly from the nominal value.

Over voltage transient protection

The TPS54231 includes overvoltage transient protection (OVTP) circuitry to minimize output voltage overshoot when recovering from output fault conditions or strong unloading transients. The OVTP circuit includes an overvoltage comparator that compares the VSENSE pin voltage with the internal threshold. When the VSENSE pin voltage is higher than 109% × Vref, the high-side MOSFET will be forced to turn off. When the VSENSE pin voltage is lower than 107% × Vref, the high-side MOSFET will be enabled again.

Thermal shutdown

When the junction temperature exceeds 175°C, the device implements an internal thermal shutdown to protect itself. Thermal shutdown forces the device to stop switching when the junction temperature exceeds the thermal trip threshold. Once the mold temperature drops below 175°C, the unit will restart the power-up sequence.

application information

Typical application diagram

step-by-step program

The following design process can be used to select component values for the TPS54231. Alternatively, Switch Pro™ software can be used to generate a complete design. Switcher™ software uses an iterative design process and accesses a comprehensive database of components as the design is generated. This section provides a simplified discussion of the design process.

To start the design process, some parameters must be determined. Designers need to know the following:

Input voltage range

The output voltage

Input ripple voltage

Output ripple voltage

Output current rating

working frequency

For this design example, use the following parameters as input parameters

On-off level

The switching frequency of the TPS54231 is fixed at 570 kHz.

Output voltage setting value

The output voltage of the TPS54231 can be adjusted externally using a resistor divider network. In the application circuit of Figure 13, this segmentation network consists of R5 and R6. The relationship of the output voltage to the resistor divider is given by Equation 4 and Equation 5:

Choose R5 to be approximately 10.0 kΩ. When using standard value resistors, slightly increasing or decreasing R5 can result in a tighter output voltage match. In this design, R4 = 10.2 kΩ and R = 3.24 kΩ, resulting in a 3.31 V output voltage. Zero ohm resistor R4 is a convenient place to disconnect the control loop for stability testing.

input capacitor

The TPS54231 requires input decoupling capacitors and, depending on the application, bulk input capacitors. A typical recommended value for decoupling capacitors is 10µF. X5R or X7R type high quality ceramics are recommended. The rated voltage should be greater than the maximum input voltage. Smaller values can be used as long as all other requirements are met; however, 10µF has been shown to work well in a variety of circuits. Also, some bulk capacitors may be required, especially if the TPS54231 circuit is not within about 2 inches of the input voltage source. The value of this capacitor is not critical, but it should be rated to handle the maximum input voltage including ripple voltage and should filter the output so that the input ripple voltage is acceptable. In this design, two 4.7µF capacitors are used for the input decoupling capacitors. They are X7R media and are rated for 50 V. Equivalent series resistance (ESR) is about 2 MΩ, and the current rating is 3 A. Additionally, a small 0.01µF capacitor is included for high frequency filtering.

This input ripple voltage can be approximated by Equation 6

where IOUT(MAX) is the maximum load current, fSW is the switching frequency, CBULK is the bulk capacitor value, and ESRMAX is the maximum series resistance of the bulk capacitor.

The maximum rms ripple current also needs to be checked. In the worst case, this can be approximated by Equation 7

In this case, the input ripple voltage will be 113mV and the rms ripple current will be 1A. It is also important to note that parasitics related to the layout and output impedance of the voltage source will greatly affect the actual input ripple. The actual input voltage ripple of this circuit is shown in the design parameters and is larger than the calculated value. This measurement is still below the specified input limit of 300 mV. The maximum voltage on the input capacitor is VIN max plus ΔVIN/2. The selected bulk and bypass capacitors are both rated at 50 V and have a ripple current capacity greater than 3 A, both providing ample margin. It is very important that the maximum voltage and current ratings are not exceeded under any circumstances.

Output Filter Components

The output filter requires the selection of two components, L1 and C2. Since the TPS54231 is an external compensation device, it can support a variety of filter component types and values.

Sensor selection

To calculate the minimum value of the output inductance, use Equation 8

KIND is a coefficient that represents the amount of inductor ripple current relative to the maximum output current. Typically, this value is up to the designer; however, the following guidelines can be used. For designs using low ESR output capacitors such as ceramic, values up to KIND=0.3 can be used. KIND=0.2 produces better results when using higher ESR output capacitors. For this design example, KIND=0.3 is used and the minimum inductance value is calculated to be 8.5μH. For this design, a large value was chosen: 10 μH.

For the output filter inductor, it is important not to exceed the rms and saturation current ratings.

In this design, the rms inductor current is 2.008 A and the peak inductor current is 2.32 A. The inductor chosen is a coiled MSS1038-103NL 10µH. Its saturation current rating of 3.04 A and rms current rating of 2.90 A meet these requirements. Smaller or larger inductor values can be used, depending on the amount of ripple current the designer wishes to allow, as long as other design requirements are met. A larger value inductor will have lower AC current and result in lower output voltage ripple, while a smaller value inductor will increase the AC current and output voltage ripple. Typically, inductor values used with the TPS54231 are in the range of 6.8µH to 47µH.

Capacitor selection

Important design factors for the output capacitor are the DC voltage rating, ripple current rating, and equivalent series resistance (ESR). The DC voltage and ripple current ratings cannot be exceeded. ESR is important because it, along with the inductor current, determines the magnitude of the output ripple voltage. The actual value of the output capacitor is not critical, but some practical limitations do exist. Consider the relationship between the closed-loop crossover frequency required for the design and the LC corner frequency of the output filter. In general, it is desirable that the closed-loop crossover frequency be less than 1/5 of the switching frequency. For high switching frequencies, such as 570 kHz for this design, the internal circuit limitations of the TPS54231 limit the practical maximum crossover frequency to about 25 kHz. In general, the closed-loop crossover frequency should be higher than the corner frequency determined by the load impedance and output capacitance. This limits the minimum capacitance value of the output filter to:

bootstrap capacitor

A bootstrap capacitor C4 is required for every TPS54231 design. The boot capacitor must be 0.1µF. The bootstrap capacitor is located between the PH pin and the bootstrap pin. The bootstrap capacitor should be a high quality ceramic type with X7R or X5R rated dielectric for temperature stability.

capture diode

The TPS54231 is designed to operate with an external capture diode between PH and GND. The diode chosen must meet the absolute maximum ratings for the application: the reverse voltage must be higher than the maximum voltage at the PH pin, which is VINMAX+0.5v. The peak current must be greater than IOUTMAX+. To improve efficiency, the forward voltage drop should be small. It should be noted that the on-time of the capture diode is usually longer than that of the high-side FET, so paying attention to the diode parameters can significantly improve the overall efficiency. Also, check that the selected equipment is capable of eliminating power losses. In this design, a diode company B240A was chosen with a reverse voltage of 40v, a forward current of 2a and a forward voltage drop of 0.5v.

Output voltage limit

Due to the internal design of the TPS54231, there are upper and lower output voltage limits for any given input voltage. The upper limit of the output voltage setting is limited by the maximum duty cycle of 91% and is given by Equation 31:

VOmax=0.91×((minimum vehicle identification number-maximum input and output×maximum output power)+VD)-(maximum input and output×maximum output power)-VD(31)

where:

VIN min=minimum input voltage

IO max = maximum load current

VD = capture diode forward voltage

RL = output inductor series resistance

This equation assumes the maximum on-resistance of the internal high-side FET.

The lower limit is limited by a minimum controllable turn-on time of up to 130ns at a junction temperature of 25°C. The approximate minimum output voltage for a given input voltage and minimum load current is given by Equation 32:

VOmin=0.096×((VIN max-IO min×Rin)+VD)-(IO min×RL)-VD(32)

where:

VIN max=maximum input voltage

IO min=minimum load current

VD = capture diode forward voltage

RL = output inductor series resistance

This equation assumes the nominal on-resistance of the high-side FET and takes into account the worst-case variation of the operating frequency set point. Any design that operates near the operating limits of the equipment should be carefully checked to ensure proper function.

Power consumption estimation

The following equations describe how to estimate the device power dissipation in continuous conduction mode. They should not be used if the device is operating in discontinuous conduction mode (DCM) or pulse skip Eco mode.

Device power consumption includes:

1) Conduction loss: Pcon=IOUT2 x RDS(on) x VOUT/VIN 2) Switching loss: Psw=0.5 x 10-9 x VIN2 x IOUT x Fsw

3) Gate charge loss: Pgc=22.8 x 10-9 x Fsw 4) Quiescent current loss: Pq=0.075 x 10-3 x VIN, where:

IOUT is the output current (A).

Rds(on) is the on-resistance of the high-side MOSFET (Ω).

VOUT is the output voltage (V).

VIN is the input voltage (V).

Fsw is the switching frequency (Hz).

so

Ptot=Pcon+Psw+Pgc+Pq

For a given TA, TJ=TA+Rth x Ptot.

For a given TJMAX=150°C, TAMAX=TJMAX–Rth x Ptot.

where:

Ptot is the total device power consumption (W).

TA is the ambient temperature (°C).

TJ is the junction temperature (°C).

Rth is the thermal resistance of the package (°C/W).

TJMAX is the maximum junction temperature (°C).

TAMAX is the maximum ambient temperature (°C).

printed circuit board layout

The VIN pin should be bypassed to ground with a low ESR ceramic bypass capacitor. Care should be taken to minimize the loop area formed by the bypass capacitor connection, the VIN pin, and the capture diode anode. A typical recommended bypass capacitor is 10µF ceramic with X5R or X7R dielectric, optimally positioned closest to the VIN pin and source of the capture diode anode. See Figure 14 for an example PCB layout. Ground D pins should be tied to the PCB ground plane of the IC pins. The power supply for the low-side MOSFET should be connected directly to the top PCB ground area for connecting the ground sides of the input and output capacitors and the anode of the capture diode. The PH pin should be connected to the cathode of the catch diode and the output inductor. Since the PH connection is the switch node, the capture diode and output inductance are very close to the PH pin, and the area of the PCB conductors should be minimized to prevent excessive capacitive coupling. In order to operate at full rated load, the top floor area must provide sufficient heat dissipation area. The TPS54231 uses a fuse lead frame, so the ground pin acts as a conductive path to dissipate heat from the die. Many applications have larger internal or back ground plane areas, and the top ground area can be connected to these areas using multiple vias under or near the device to aid heat dissipation. Other external components can be placed roughly as shown. Acceptable performance may be obtained using an alternative layout scheme, but this layout has been shown to produce good results and serves as a guideline.

UV Radiation Resistor Separator

Estimating circuit area

The estimated printed circuit board area of the components used is 0.68 square inches. This area does not include test points or connectors.

Electromagnetic Interference (EMI) Considerations

As EMI becomes a concern in more and more applications, the internal design of the TPS54231 takes measures to reduce EMI. The high-side MOSFET gate drive is designed to reduce PH pin voltage ringing. Inter-IC tracks are isolated to reduce noise sensitivity. In order to reduce parasitic effects, the wrapped junction scheme is adopted.

For optimum EMI performance, the selection of external components is as important as the layout of the circuit board. Follow the step-by-step design procedure above to prevent potential EMI issues.